EMC FLEX BLOG A site dedicated to Automotive EMC Testing for Electronic Modules

CISPR 25 Conducted Emissions Measurements.

  CISPR-25 indicates that both CE-V and CE-I must be carried out to validate an automotive electronic product.


CISPR-25 indicates that both CE-V and CE-I must be carried out to validate an automotive electronic device.

CE-V in dBuV is measured on B+ and GND lines using the LISN port.

CE-I in dBuA is measured using a “current probe” clamped at 5 cm, then at 75 cm from DUT’s connector. The probe is clamped on the whole harness, then on each connector separately. The RF noise measured may be coupled from DUT directly as well as from wire-to-wire along the 1.7 m test harness.

CISPR 25 is not very specific about supply lines CE “redundancy”, therefore we test everything for CE-I.

Chrysler is the only OEM that specifies in CS.00054 as exception from CISPR 25 to remove from “current probe” all Supply Lines (power and ground).

CS.00054 is asking to run CE-I on all wires not tested at CE-V, however measurements are aquired only at 5 cm from DUT's connector.



Christian Rosu

CISPR 25 Ground Plane Size

  Differential-mode RF emissions in a CISPR 25 component level configuration occur due to


Differential-mode RF emissions in a CISPR 25 component level configuration occur due to the flow of current (IDM) via signal paths in which the forward and return conductors are not routed together, thereby forming a conductor loop. The resulting magnetic field from the conductor loop is proportional to the current IDM, the area of the loop and the square of the frequency of the RFI current.

Common-mode RF emissions occur due to undesired parasitic effects, e.g. due to inductances in the current return path or unsymmetries during signal transmission. If we connect a cable to a DUT of it may function like an antenna allowing a common-mode current ICM to flow. Both signal and power supply lines can function as efficient antennas. Here, our rule of thumb is that line lengths that do not exceed λ/10 are uncritical, whereas longer lines (e.g. λ/6) must be treated as potential sources of RF emissions.

The magnitude of the voltage drop on the ground plane and thus the magnitude of the common-mode current coupled into the connected line are determined by the parasitic inductance and the slope steepness of the signal.





We cannot assume that differential mode radiated emissions are not dominant nor an infinite ground plane. A ground plane with finite width has inductance.

Common-mode RF emissions can also occur due to differential mode signal transmission.
If the parasitic terminating impedances of a differential mode transmission path differ substantially, in addition to the desired differential-mode current IDM a common-mode current ICM will also flow via the ground plane that connects the transmitter and receiver modules. This unwanted ground current ICM can then also be coupled into lines connected to DUT and cause emissions in the far field.

The strength of the common mode current and the level of radiated emissions depend on the inductance of the ground plane. The value of this inductance depends on the structure of the transmission line.

The ground plane inductance in a symmetric structure is:
L21 = (µ0/) * ln((/W)+1)
W is the width of the ground plane
t is the height of the harness

The ratio of the height of the harness and the width of the ground plane determines the GP inductance.



As the harness is closer to the edge of the ground plane, the measurement tolerances are higher since the ground plane inductance increases. The tolerances in RE measurments are acceptable when the distance of the harness to the ground plane edge is 10 cm.
Since common mode radiated emissions occur through the ground plane (or the whole setup), the length of the ground plane can impact the tolerances in RE measurments. Longer the ground plane, higher the radiated emissions level.


Christian Rosu, 2022-03-07



RF Boundary in automotive EMC for electronic components

RF Boundary is the element of an EMC test setup that determines what part of the harness and/or&nbsp

RF Boundary is the element of an EMC test setup that determines what part of the harness and/or peripherals is included in the RF environment and what is excluded. It may consist of, for example, ANs, BANs, filter feed-through pins, RF absorber coated wire and/or RF shielding.


RF Boundary is also an RF-test-system implementation within which circulating RF currents are confined


  • to the intended path between the DUT port(s) under test and the RF-generator output port, in the case of immunity measurements (ISO 11452-2, ISO 11452-4, ISO 1145-9), and
  • to the intended path between the DUT port(s) under test and the measuring apparatus input port, in the case of emissions measurement (CISPR 25),


and outside of which stray RF fields are minimized.


The boundary is maintained by insertion of BANs, shielded enclosures, and/or decoupling or filter circuits. The ideal RF boundary replicates the circuitry of the device connected to DUT in vehicle.

The standard test harness lenght for automotive EMC electronic components is (1700mm -0mm / +300mm). This 1.7m test harness runs between the DUT and the Load Simulator (Shielded Enclosure) that plays the role of RF Boundary.


If the Load Simulator enclosure does not include all DUT loads and activation/monitoring support equipment, additional support devices may be placed directly on the ground plane. The connection of additional devices to LS enclosure must be done via short wiring running on the ground plane.


Testing at subsystem level is preferable to any simulation. Whenever possible, use production intent representative loads.


Running long coax cables directly from DUT outside the chamber via SMA bulk filter panel would violate the 1.7m test harness length rule invalidating the test result. Ideally is to use Fiber Optic to exchange data with devices placed outside the test chamber.


Running long coax cables between Load Simulator and a support device placed outside the chamber is acceptable as long as the I/O line in question is not just an extension from DUT without proper RF boundary at the end of maximum 2-meter length of standard test harness.


It is critical to use the test harness length as defined by CISPR-25, ISO 11452-2, ISO 11452-4, and ISO 11452-9 to achieve valid compliance for your product. The length of the test harness as well as the grounding method (remote vs local) can result in different RF emissions level. Longer the test harness, higher RF emissions above 100 MHz due to its resonance pattern. The local grounding would show less magnitude variation across resonance peaks above 100MHz.


Christian Rosu



Baterry Line Transient Pulse 1b

24. January 2022 10:47 by Christian in EMC/EMI, EMC TEST PLAN, OEM Specs, Test Methods
Pulse “1b” is defined differently by various international standards and/or OEM EMC spec

Pulse “1b” is defined differently by various international standards and/or OEM EMC specs.

Daimler and Chrysler have quite a similar definition for Pulse 1b. Us = 30V.


Nissan requirement for Pulse 1b is quite different (-100V).

The old 2007 version of SAE J113-11 is also significantly different for multiple pulse parmeters.

Christian Rosu, 2022-01-24


18. November 2021 18:28 by Christian in EMC/EMI, EMC TEST PLAN, Grounding, PCB, Shielding
Near-field interference (crosstalk) is a major issue in electronic devices and systems when comes to EMC compliance.

Near-field interference (crosstalk) is a major issue in electronic devices and systems when comes to EMC compliance. To reduce crosstalk, as well as far-field interference, the transmission lines can be shielded. The length of transmission lines and spacing between the conductors should be as small as possible. The steel tube around the untwisted pair is superior to both the aluminum and copper tube due to its magnetic properties.


 Untwisted Pair

The length of the conductors and the spacing between them must be short.

Untwisted Pair Inside Copper Tube

Less susceptible to near filed magnetic noise. Copper is non-magnetic with μr (relative permeability) = 1. A very small induced bucking current in the Cu tube will generate a small  counter magnetic field.

Untwisted Pair Inside Grounded Aluminum Tube

Less susceptible to near filed magnetic noise. Aluminum is non-magnetic, 0.61 of copper conductivity. Therefore, the magnitude of the counter magnetic field is less. The grounded Al tube reduces near-field electric emissions susceptibility and static charge buildup on the shield being less expensive.

Untwisted Pair Inside Steel Tube

The steel tube & untwisted pair is better than Al or Cu tube due to its magnetic properties (μr = 1000 @ low frequencies):
(i) it increases the absorption of the magnetic fields
(ii) redirects the magnetic fields away from the tube's interior

 Twisted Pair

Twisted pair without any shielding is ranked higher than the untwisted pair in a steel tube. Per Lenz's law, the magnetic field will induce a voltage in a loop of wire. The orientation of the loop affects the sign of this voltage. Twisting the two wires forces the induced voltage in neighboring loops to be of opposite polarity. By summing all of the induced voltages from each of the loops generated by the twisting, the net induced noise voltage is significantly less than without the twisting. The sum is theoretically zero for an even number of loops. Twisting the wire is probably one of the least expensive methods to decrease the susceptibility of a cable to magnetic fields.

Twisted Pair Inside Steel Tube

Twisted pair inside a steel tube is the least susceptible to magnetic fields. The steel tube absorbs and redirects the magnetic fields. Steel is relatively inexpensive, but it is heavy. To avoid rusting it should be galvanized.




  •  The shielded twisted pair with both the source and load grounded offers only 2 dB less susceptibility to low-frequency noise. Although the grounding of the shield will reduce electric field emissions, many of the advantages of using twisted pair are lost since the load and source are not balanced (load & source are both single-ended grounded).
    The return current path is divided between the return conductor of the twisted pair and the ground plane.
    At low frequencies, a majority of the current tends to return via the low-impedance ground plane. At higher frequencies, the current tends to return along a path nearest to the forward signal current - the twisted pair conductor.


  •  The 'least immune, or most susceptible, cable of those listed to low-frequency noise (both electric & magnetic) is the coaxial structure where both the source and load are grounded and the shield is only connected to the source ground. Although the grounding of the shield will reduce electric field emissions from the center conductor, the outer shield has virtually no influence on the magnetic field susceptibility.


  •  The return current for both the signal and noise must be via the ground path between the load and source. This current-path loop can be large. The effective area for either magnetic emissions or magnetic pickup can therefore also be large. This dosed current path through the ground is referred to as a ground loop.


  •  A significant improvement in performance is obtained when a twisted pair is used and the load is balanced.
    The load is floating with neither end of the load connected to ground.
    At low frequencies, the parasitic coupling between the load and nearby grounds is small, and the signal current should mostly return via the return conductor of the twisted pair. The effective pickup area of the complete current path is small, and the twisting produces alternating polarity induced voltages in each of the loops. Since there is no surrounding shield, there is no electric field shielding. However, if the line is balanced, the capacitive coupling (i.e., near-field electric coupling) to each line should be about the same, and the net electric-field induced noise across the load should be negligible.


  •  When the shield or outer conductor of the coaxial cable is grounded at both ends (source & load), the return current will divide between the shield and ground plane. If the frequency of the signal is much greater than the cutoff frequency of the shield, then most of the current will return via the shield. If the frequency of the signal is much less than the cutoff frequency of the shield, then most of the current will return via the ground plane. This scenario is slightly better than the shielded twisted pair arrangement shown above dur to lower resistance of the shield relative to the return conductor of the twisted pair. Noise current passes through the shield and returns through the ground path. Most of the signal current, not shown here, returns via the return of the twisted pair conductor. A potential disadvantage of multiple ground points is that noise currents can exist along the shield. In addition, since the shield has a nonzero impedance, the noise voltage can also vary along the shield. Since capacitive coupling exists between the shield and each of the twisted pair conductors, noise will be induced across the load unless the line is perfectly balanced. The capacitive coupling from each conductor to the shield should be nearly the same. This noise will be a function of the distance along the line.


  •  The symmetry of the concentric shield is utilized when the load is floating and the shield is connected to the load. Furthermore, the voltage at the load end of the shield is closer to the voltages along the twisted pair conductors at the load. The currents from the shield to the twisted pair conductors via the parasitic capacitance are less since the voltage across the parasitic capacitance is smaller. DISAVANTAGE: any noise current that couples into the system can now exist on the shield and signal return. The conservative approach is to avoid noise currents on the signal and return conductors even when the system is (partially) balanced.


Christian Rosu, Nov 18, 2021.